Microwave vacuum window having wide bandwidth

ABSTRACT

A distributed microwave window (12) couples microwave power in the HE11 mode between a first large diameter waveguide (32) and a second large diameter waveguide (34), while providing a physical barrier between the two waveguides, without the need for any transitions to other shapes or diameters. The window comprises a stack of alternating dielectric (14) and hollow metallic (16) strips, brazed together to form a vacuum barrier. The vacuum barrier is either transverse to or tilted with respect to the waveguide axis. The strips are oriented to be perpendicular to the transverse electric field of the incident microwave power. The metallic strips are tapered on both sides of the vacuum barrier, which taper serves to funnel the incident microwave power through the dielectric strips (14). A suitable coolant flows through a coolant channel (18) that passes through the metallic strips (16). The microwave window further includes an impedance matching transition (15) between the tapered metal vanes and insulating dielectric material used to create the vacuum barrier of the window. Such impedance matching transition comprises one or more quarter wave (λ/4) matching sections in the individual vane structure that achieves the required impedance match. The effect of such impedance match is to render the dielectric material, e.g., sapphire, non resonant. Such non-resonance significantly widens the bandwidth of the window.

This application claims the benefit of U.S. Provisional application Ser.No. 60/001,208, filed Jul. 18, 1995.

BACKGROUND OF THE INVENTION

The present invention relates to large diameter microwave waveguides,and more particularly to a distributed window that may be used in suchwaveguides to couple high frequency, high power microwave radiationthrough a vacuum barrier within the waveguide without overheating,significant mode conversion, or reflection of incident power. Even moreparticularly, the invention relates to the use of an impedancetransition built into the vanes of the vacuum barrier to increase thebandwidth of the window. Such impedance transition, which comprises oneor more quarter wave matching sections in the individual vane structure,makes the dielectric material used as part of the vane structure nonresonant. In turn, this non resonant condition reduces the powerdissipated in the dielectric, and thereby increases the power handlingability of the window.

A waveguide window in a microwave power system permits power to becoupled from a first waveguide to a second waveguide, but presents aphysical barrier between the two waveguides. The physical barrier allowsthe waveguides to contain different gases or to be at differentpressures, and one or both waveguides may be evacuated. For example, inhigh power microwave vacuum devices, such as gyrotrons and the like, theoutput power must be coupled between an evacuated chamber or waveguidein the gyrotron device, through one or more waveguide windows, into awaveguide having a gaseous environment. The one or more waveguidewindows thus provide a hermetic seal between the two media. Also, infusion reactors where microwave power is added to a plasma, the physicalbarrier of a microwave window may be placed near the reactor to confinethe constituents of the plasma.

One type of microwave window known in the art is a double-disk window. Adouble-disk window can be tuned over a limited frequency range tocompensate for errors in window thickness or unit-to-unit variation ingyrotron output frequency.

Another type of microwave window known in the art is described in U.S.Pat. No. 5,061,912, incorporated herein by reference. The type ofmicrowave window disclosed in the '912 patent is a distributed windowthat forms part of a phase velocity coupler. The type of couplingprovided by the described window is between two identical corrugatedrectangular waveguides, each of which is many (e.g., >15) free spacewavelengths, λ₀, wide in one transverse dimension but only 2 to 3 λ₀ inthe other dimension. A transition from circular corrugated waveguidemany λ₀ in diameter propagating the HE₁₁ mode, which is a preferredmethod of low loss transmission for high power millimeter wavelengthmicrowaves, to this rectangular corrugated waveguide, can always bemade. However, if the circular waveguide is very large, e.g., 30λ₀ indiameter, many modes which can propagate in the larger circularwaveguide are cut off in the rectangular waveguide. Although ideally,only one mode is emitted from the source, typically a gyrotron, andpropagated through the system, in reality there is often a few percentof other modes present, which might be reflected back to the source withdeleterious effects by such a transition. Hence, there is a need in theart for a microwave window that can be used to directly and efficientlycouple high frequency microwave power between two large diameterwaveguides without the need for any transitions to other shapes andsizes.

Such needs are met, at least in part, by the large diameter distributedmicrowave windows disclosed in applicant's prior U.S. Pat. No. 5,313,179and 5,400,004. Advantageously, such large diameter microwave windows areparticularly well-suited for use with the new generation of gyrotrons,such as the Russian 500 kw, 110 and 140 GHz gyrotrons which have theHE₁₁ output mode, which are most compatible with a large outputdiameter. Unfortunately, however, depending upon the type of dielectricmaterial that is used in the vane structure of such windows, thebandwidth of such distributed windows may be very narrow. For example,when sapphire is used as the dielectric, the high dielectric constant ofsapphire renders the bandwidth of the window very narrow. Such narrowbandwidth may cause problems with the gyrotron operation, and could makethe distributed window less attractive than the double-disk window.Hence, there is a need in the art for a way to widen the bandwidth ofthe distributed microwave window.

SUMMARY OF THE INVENTION

The present invention addresses the above and other needs by providing adistributed microwave window similar in construction to the windowsdescribed in the above-referenced '179 and '004 patents, but whichfurther includes an impedance matching transition between the taperedmetal vanes and insulating dielectric material used to create the vacuumbarrier of the window. Such impedance matching transition comprises oneor more quarter wave matching sections in the individual vane structurethat achieves the required impedance match. The effect of such impedancematch is to render the dielectric material, e.g., sapphire, nonresonant. Such non-resonance significantly widens the bandwidth of thewindow.

In accordance with one aspect of the invention, once the requiredimpedance match is achieved, a significant reduction in the powerdissipated in the dielectric material also results. If the dielectricmaterial is sapphire, for example, the reduction in power dissipated inthe sapphire is almost 50%. Such reduction in power dissipation meansthat the power handling ability of the window is greatly increased.

As described in the referenced '170 and/or '004 patents, the basicdistributed microwave window of the present invention includes a barrierformed from a stack of alternating dielectric and hollow metallicstrips, brazed together to make good thermal contact with each other andto form a vacuum seal. The hollow metallic strips are positioned to beperpendicular to the transverse electric field of the incident wave. Themetallic strips further include a specified taper that deflects theincident microwave power away from the metallic strips and through thedielectric strips. A coolant is pumped through the hollow metallicstrips in order to remove heat generated at the dielectric strips by themicrowave power passing therethrough.

The impedance matching transition in accordance with the presentinvention includes one or more quarter wave matching sections thatinterface the tapered metallic strips with the dielectric strips.

BRIEF DESCRIPTION OF THE DRAWINGS

The above and other aspects, features and advantages of the presentinvention will be more apparent from the following more particulardescription thereof, presented in conjunction with the followingdrawings wherein:

FIG. 1 shows a distributed window made in accordance with the presentinvention that couples two large diameter waveguides;

FIG. 2A shows a typical cross-sectional view of a portion of a barrierused to form the microwave window in accordance with the inventiondisclosed in the '179 and '004 patents;

FIG. 2B illustrates a cross-sectional view through one of the coolantchannels of a metallic strip used within the microwave window of FIG.2A;

FIG. 3 depicts a cross-sectional view as in FIG. 2B where the barriercreated by the stacked alternating dielectric and metallic strips istilted relative to the waveguide axis;

FIG. 4 diagrammatically defines the dimensions used in a thermalanalysis of the invention;

FIG. 5 defines the coordinate system and linear dimensions associatedwith an ohmic loss analysis of the invention;

FIG. 6 shows a typical cross-sectional view of a portion of a barrier asin FIG. 1 with blunt tapers;

FIG. 7 illustrates a cross-sectional view of a portion of a barrier usedto form the microwave window as in FIG. 2A, and further shows the use ofcorrugations on opposing surfaces of the dielectric strips in order tolower the effective dielectric constant of the dielectric strips, andthereby minimize the dielectric and ohmic losses through the barrier;

FIG. 8 depicts a frontal view of a portion of the barrier of FIG. 7, andshows the preferred orientation of the corrugations relative to thedielectric and cooling strips;

FIG. 9 is a side sectional view taken along the line 9--9 of FIG. 8, andillustrates the parameters used to define the corrugations;

FIG. 10 shows a transmission line model useful in explaining andunderstanding the present invention;

FIG. 11 illustrates a cross-sectional view of a portion of the barrierused with the window of the present invention, and illustrates one typeof quarter wave impedance matching section that may be used with theinvention;

FIG. 12 shows an enlarged view of a portion of the barrier of FIG. 11;

FIG. 13A depicts a cross-sectional view of a first alternativeembodiment of the barrier used with the window of the present invention,illustrating the use of a series of quarter wave impedance matchingsections between the tapered vanes and the dielectric;

FIG. 13B depicts a cross-sectional view of a second alternativeembodiment of the barrier used with the window, illustrating the use ofan identical series of grooves (functioning as impedance matchingsections) between the tapered vanes and the dielectric;

FIG. 14 shows an equivalent transmission line circuit useful inanalyzing the performance of the microwave window of the invention;

FIG. 15 is a graph illustrating power transmitted to and reflected fromthe load through the window as a function of frequency;

FIG. 16 depicts the equivalent circuit of the window on either side ofthe dielectric;

FIG. 17 shows an equivalent circuit of the window configured in terms ofvoltages and characteristic admittances;

FIG. 18 is a graph that illustrates the ratio of reflected to forwardpower, and the ratio of power delivered to the load and power generatedas a function of frequency for the waveguide window structure of FIG.13A;

FIG. 19 shows additional detail associated with the second alternativeembodiment of FIG. 13B;

FIG. 20 is the equivalent transmission line circuit for the waveguidestructure shown in FIG. 19; and

FIG. 21 is a graph that illustrates the ratio of reflected to forwardpower, and the ratio of power delivered to the load and power generatedas a function of frequency for the waveguide window structure of FIG.13B.

Corresponding reference characters indicate corresponding componentsthroughout the several views of the drawings.

DETAILED DESCRIPTION OF THE INVENTION

The following description is of the best mode presently contemplated forcarrying out the invention. This description is not to be taken in alimiting sense, but is made merely for the purpose of describing thegeneral principles of the invention.

FIGS. 1-9 are described in applicant's prior U.S. Pat. No. 5,400,004,incorporated herein by reference. It is noted that the '004 patent is acontinuation-in-part of applicant's prior U.S. Pat. No. 5,313,179, andthat the entire substantive disclosure of the '179 patent is included inthe '004 patent.

FIGS. 1-9 describe a distributed microwave window that is suitable forsome applications, e.g., as described in the previously referenced '179or '004 patents. However, it has recently been discovered that thebandwidth of the distributed window as disclosed in the '179 or '004patents (hereafter the "179/004 window") is really rather narrow.Although its bandwidth is accurately represented by that of a simplesingle disk sapphire window of the same thickness as that used in the179/004 window, it turns out that such bandwidth is rather narrow forsome applications. It had been supposed that the bandwidth of the179/004 window would be better than the bandwidth of, e.g., Varian'sdouble disk window. In one sense, the bandwidth of the 179/004 windowmay be better than that of the Varian window. However, the Varian windowcan be tuned by varying the spacing between disks, whereas the179/004window cannot be tuned. Hence, there is the probability that theoperation of a gryroton could be adversely influenced by using the179/004 window. In particular, a reflection from a 179/400 window couldraise the cavity "Q", thereby increasing cavity dissipation, or reducingefficiency.

A method is described in the '004 patent for transverse grooving of theedge of the sapphire (dielectric), where the grooves were to act as amatching section of intermediate dielectric constant. To date, thisscheme has never been implemented to applicant's knowledge because therequired grooves, which must be spaced less than 1/2 wavelength in thedielectric, appear to be too fine to make economically. Making finegroves of this type is particularly problematic since the strength ofsapphire depends critically on surface finish, which is difficult tocontrol in the grooves. There also appears to be some uncertaintyregarding the power dissipation in the grooved sapphire section.

In a plane disk window, the use of a 1/4 wavelength (λ/4) matching layeron each surface, either by grooving or by application of a layer of adifferent (intermediate) dielectric constant, is the only known way tomake the window broadband.

In the case of the distributed window (i.e., the 179/004 window),however, in which the power is divided among many narrow slot windows,the spacing of which is less than one free space wavelength, it ispossible to utilize the metal structure supporting the strips as amatching section, since only one transverse mode is supported in theslots in which the dielectric strip windows are located.

The result of having such matching sections is to widen the bandwidth ofthe window, and thereby reduce the dissipation in the dielectric vacuumbarrier. Both effects are due to the absence of a standing wave in thedielectric by the use of matching sections. A standing wave does existin the matching sections, but because they are much shorter and containno dielectric, the bandwidth is greatly increased and lossess reduced.

For comparison, reference should be made to FIG. 2A which shows across-sectional view of the conventional (original) distributed windowsection as taught in the '179 patent. The window includes a barrier 12made up of vanes comprising alternating tapered metallic strips 16,having coolant channels 18 therein, and dielectric strips 14. Withoutthe dielectric strips 14, the structure is very wide band.

Referring to the transmission line model of FIG. 10, it is well known ifZ₀ is the system impedance, and if Z₂ is the impedance of anothersection of the transmission line, that at a given frequency f₀, if λ₀=c/f₀, where c is the velocity of light in the medium, then a matchingsection of impedance ##EQU1## and length λ₀ /4, where λ₀ corresponds tothe wavelength of the center frequency, f₀, of the desired band, willgive the best match between lines of impedance Z₀ and Z₂. Thetransmission line model of the matched window would then appear asabove, in the simplest case.

Turning to FIG. 11, there is shown one embodiment of a vane structurethat may be used to match impedances in accordance with the presentinvention. As seen in FIG. 11, the tapered metallic strips 16, areseparated by a dielectric window 14, with a quarter wavelength, λ/4,matching section 15 on each side of the dielectric window 14. Ingeneral, the dielectric window is made of a strip of sapphire having aheight b, and a width nλ/(2√ε). The impedance Z of a waveguide sectionis proportional to b/√ε, where ε is the dielectic constant of anydielectric filling the waveguide section. As an example, since ε=9.4 forsapphire, and the height of the sapphire strip is, e.g., 0.022 inches(0.0559 cm), the vacuum matching section height w must be on the orderof ##EQU2## This is a very small gap, and, as seen in FIG. 11, it coverspart of the sapphire.

To work properly, the overlapping edge should be brazed to the sapphire.For the structure shown in FIG. 11, it may be difficult to be certain ofthis braze, and any overflow would be difficult to clean. Further, asseen best in the enlarged view of the sapphire/metal junction or jointin FIG. 12, it could be very difficult to inspect the joint for gaps.

Alternative embodiments to the structure shown in FIGS. 11 and 12 areshown in FIGS. 13A and 13B. Such embodiments, rather than using a lowimpedance matching section, follow the sapphire with a at least one λ/4section of impedance Z₀ (vacuum section of same height as sapphire).Referring first to FIG. 13A (FIG. 13B is described more fully below),since the sapphire impedance is Z₀ /√ε, the impedance at the oppositeend of a λ/4 matching section is √ε·Z₀ at the design frequency. To matchthis to Z₀ requires another λ/4 section of impedance ε^(1/4) ·Z₀, whichalso requires a waveguide height (gap size) greater than the sapphireheight, so there is a clear line of sight to the dielectric strip(having a width b') without covering or blocking any portion of thedielectric strip, i.e., so that there is a clear aperture of width b' tothe sapphire.

As described above, it is thus seen that for the waveguide windowembodiment shown in FIG. 13A, an impedance matching section 15' is usedwhich is made up of two sections of size λ/4. Thus, as seen in FIG. 13A,the second and third set of opposing sides of the metallic strip 16(which has generally a hexagonal-shaped cross section) combine to form ataper 22 on each side of the vacuum barrier 12 for each one of saidmetallic strips 16 that forms part of the microwave window barrier 12.Each of the tapers 22 has a tip or ridge 26 that extends the length ofthe metallic strip. The ridge is a distance L from the beginning of theimpedance matching section 15'. The impedance matching section 15', inturn, has a length of λ/4+λ/4 or λ/2, where λ is the free spacewavelength of the electromagnetic radiation propagating through thewaveguide. Thus, where the dielectric strip 14 has a thickness d, it isseen that the overall width of the barrier 12 of the embodiment shown inFIG. 13A, from tip-to-tip, is 2(L+λ/2)+d.

To better understand the benefits of the arrangement shown in FIG. 13A,it is helpful to analyze the equivalent transmission line circuit shownin FIG. 14. From FIG. 14, the following relationships between thecurrent and voltage at each node along the waveguide (transmission line)may be established: ##EQU3##

Starting from the matched condition on the left side of FIG. 14, it isseen that V₁ =I₁ Z₀. One can find V₂,I₂ from Eq. (1a) above, V₃,I₃ fromEq. (1b) above, etc., to get V⁺ ₆ and V⁻ ₆ from Eq. (1f). The power fromthe generator is then ##EQU4## and the power to the load is ##EQU5## thepower reflected back to the generator is ##EQU6## Since V₁ is given andarbitrary, renormalization can be done so that the incident power,P_(inc), from the generator is 1, the reflected power P_(refl) from theload is ##EQU7## and the power transmitted to the load P_(trans) is##EQU8## In this analysis, and with reference to FIG. 14, it is notedthat l=λ/4, where λ is the free space wavelength at the design centerfrequency, ##EQU9## where n is an integer, ##EQU10## wavelength in thedielectric of dielectric constant ε, and the λ's represent complexpropagation constants. That is, γ₁ =α₁ +iβ₁, where ##EQU11## Similarly,γ₂ =α₂ +iβ₂, where α₂ =R_(VAC) /b_(O) and β₂ =k₀ ; and γ₃ =α₃ +iβ₃,where ##EQU12## Here, k₀ =2πf/c, where f is the applied frequency and cis the freespace velocity of light, b₀ is the height of the dielectric,R_(VAC) is the surface resistance of the frame material normalized to377 ohms, and R_(E) is the surface resistance of the sapphire brazematerial seen at the edge of the sapphire, normalized to 377 ohms.

By way of example, if b₀ =0.022 inches, ε=9.4 for sapphire, R_(VAC)=0.26 ohms at 170 GHz, and R_(E) =0.52 ohms at 170 GHz. Note that R_(E)is multiplied by √ε in the term α₃ since the impedance is ##EQU13## Theadditional dielectric loss is not specifically included, but can beconsidered to be lumped into R_(E). In this example n=3, (3-λ/2 in thedielectric).

FIG. 15 is a graph that shows the dramatic reduction in reflection awayfrom the design center frequency when the matching sections are used, asdescribed above, compared to a window without matching sections. Withmatching sections, the reflection never exceeds 6%, and that occurs 10GHz away from the center frequency of 170 GHz. Without the matchingsections, a 6% reflection occurs 3.5 GHz away from 170 GHz, and thereflection keeps increasing to a maximum of about 60%. Such strongreflections could affect the gyrotron operation if the window centerfrequency deviates from the gyrotron operating frequency. For a 2%reflection, with matching sections, the bandwidth is over 8 GHz (over 4GHz on either side of the center frequency 170 GHz). In contrast, withthe unmatched window of the same thickness, the 2% bandwidth is <4 GHz.Note that this occurs with a sapphire window only 1 h wavelengths (inthe sapphire) thick, or 0.034 inches at 170 GHz. A thicker windowwithout matching sections would be proportionally narrower in bandwidth.

As also seen in the graph of FIG. 15, less dramatic, but of at leastequal importance, is the reduction in loss at the center frequency whenthe window is matched. This results in only a traveling wave passingthrough the dielectric, compared to the prior art distributed (or singleor double disk) resonant window, where reflection is avoided by makingthe window an integral number of 1/2 wavelengths (in the dielectric)thick, thereby resulting in a standing wave in the dielectric, whichincreases both dielectric loss and ohmic dissipation at the walls of thewaveguide. The ratio of dissipation for the unmatched resonant windowcompared to that of the matched window with a traveling wave is(ε'+1)/(2√ε'), or about 1.70 for ε'=9.4 (note, ε' is the real part ofε), appropriate for sapphire. If the losses in the matching section areincluded, as is true for the solid curves in FIG. 15, the reduction intotal loss will be less than the case in which the matching sections arelossless.

However, it is pointed out that the total loss is not the most importantconsideration. The maximum continuous wave (CW) power that the windowcan handle is limited by the Watts/cm² at the sides of the sapphirewhere they are brazed to the metal frame, the back sides of which arewater cooled. If the combined loss from dielectric heating and ohmicloss at the sapphire-braze interface can be reduced by 1/1.7, the windowwill be able to pass 1.7 times as much power compared to the unmatched,resonant window, even though the dissipation in other parts of the frameis increased. This is because the matching sections have a larger areathat is water cooled than the sapphire, and the loss in the matchingsections is still not nearly as large as the total loss in the (matched)sapphire window. As a result, it is still the Watts/cm² at thewindow-frame interface that limits the power handling at the window.

The model circuit shown in FIG. 14 does not include the effect of thestep discontinuity, which, as presented in the Waveguide Handbook by N.Marcuvitz (published by Dover books, NY, N.Y.), pages 307-309, has theeffect of introducing an equivalent shunt capacitance at the step. Theequivalent circuit on either side of the dielectric is then as shown inFIG. 16. The circuit shown in FIG. 16 is written in terms of admittancesrather than impedances to simplify treating the shunt elements. Thus, inFIG. 16, G is the (real) admittance seen at the interface with thedielectric, assuming a traveling wave (G/Y₀ '=√ε), Y₀ and Y₀ are thecharacteristic admittances of the parallel plate waveguide sections ofheights b' and b, respectively, to use the notation of Marcuvitz, andthe C's are the capacitances due to the discontinuity, each having anadmittance iB, which is purely imaginary. Because of these capacitances,the optimum spacing between steps is no longer λ/4, but will be denotedby l. Since the dielectric-vacuum interface does not introduce such acapacitance, because there are no higher modes excited at such aninterface, no correction is needed in the distance from the dielectricto the first step.

The conductance G, transformed by the right hand λ/4 section,contributes conductance ##EQU14## to Y.sub.α, so the total admittanceY.sub.α is Y.sub.α =S₁ +iB. The ratio of the admittance Y_(b) /Y₀ may beexpressed as ##EQU15## The objective is to transform S₁ to another realadmittance S₂ at Y₆. For a single section transformer, S₂ would be Y₀ ',while if there is a following section to the left, S₂ would be anintermediate value between S₁ and Y₀ '. In any event, Y₆ =S₂ is to bereal. In addition, since for B=0, βl₀ =π/2, let βl=π/2+δ for B>0. Thentanβl=-1/tanδ.tbd.-1/Δ. Then defining S₁ .tbd.S₁ /Y₀ ', B.tbd.B/Y₀ ' andα.tbd.Y₀ /Y₀ '=b'/b, where b' and b are the respective heights of the Y₀' and Y₀ admittance waveguides, it is seen that ##EQU16## The real andimaginary parts of Eq. (2) give Δ=-B/α and ##EQU17## respectively. SinceB is ≧0, and α is positive, Δ≦0 which means l≦λ/4. Eliminating α fromthe two equations, it is seen that ##EQU18## Thus, it is seen than##EQU19## Since β B depends on α in the analysis in Marcuvitz, it isnecessary to proceed iteratively, starting with B=0, giving ##EQU20##and using the value of B so obtained to give Δ. This approach ispractical for small B (e.g.,B² <<S₁ S₂), since α then depends onlyweakly on B, while Δ has a first order dependence on B.

For the preceding example, with ε=9.4, and b'=0.022", it is seen thatb=0.022·ε^(1/4) =0.0385 inches; and according to Marcuvitz, ##EQU21##Since the other transverse dimension is much larger than a free spacewavelength, λ_(g) ≈λ₀ =c/f, it is seen that tanδ=-0.433, and the ratioof the corrected transformer section length l to the uncorrected lengthl₀ =λ/4 is l/l₀ =0.74.

To determine the reflection and transmission properties of the windowwith the corrected transformer section length, an analysis similar tothat presented above in connection with FIG. 14 is performed. Inparticular, reference is made to FIG. 17, where it is seen that l₁ =l₅=l, the corrected length, l₂ =l₄ =λ/4, and l₃ =nλ/(2√ε. If α is setequal to α.tbd.Y₀ /Y₀ '=b'/b, then, from right to left in FIG. 17, witha termination G=Y₀ ' on the right, it is seen that ##EQU22##

In the above equations, V⁺ e⁺γZ represents a wave traveling to the rightwith propagation constant γ=α+iβ, with α comprising the attenuationconstant, and β=ω/c comprising the phase shift/unit length in the zdirection, except in the dielectric, where ##EQU23## Likewise, V⁻ e⁻γZrepresents a wave traveling to the left. B is the (positive) capacitivesusceptance of the discontinuity capacitance which, because of symmetry,is the same at all the steps.

Continuing with the analysis of FIG. 17, it can be shown that: ##EQU24##Finally, ##EQU25## where Y₀ ' is assumed to be the characteristicadmittance of the input (and output) waveguides. The ratio of reflectedforward power at the input to the circuit shown in FIG. 17 is just |V⁻ ₆/V⁺ ₆ |², the ratio of reverse to forward power flowing through thedielectric is just |V⁻ ₃ /V⁺ ₆ |², while the ratio of generator forwardpower incident on the circuit from the left to the power in the load isjust |V₁ /V.sub.⁺ ₆ |². These expressions may be evaluated numericallyin sequence, starting with an arbitrary value of V₁ and Y₀ ', since onlythe ratio of voltages and admittances are important in the final result.

Using the same numerical example presented previously, and assuming thevalue of B/Y₀ '=0.227 and l₁ (=l₅)/(λ/4)=l/l₀ =0.74 as used previously,it is seen that the reflection, transmission, and reflection coefficientin the dielectric are as shown in the graph of FIG. 18. Correcting forthe discontinuity capacitance actually widens the bandwidth, or at leastreduces the peak reflections compared to the uncorrected examplepreviously presented (see FIG. 15), which uses a different verticalscale. The curves shown in FIG. 18 assume the same α's as used in FIG.15, R_(VAC) =0.26 ohms, R_(dielectric) =R_(E) =0.52 ohms.

Although the result presented in FIG. 18 is very attractive, it ispossible that a further correction may be required in the length of thematching section to account for any interaction that occurs between thesteps due to evanescent higher mode fields excited by such large stepsso close together, and further due to the waveguide height b of thetransformer section being larger than λ/2. (Note, for the examples givenat 110 GHz, λ/2=0.34 inches, while b=0.0385 inches.) The next highermode in fact can propagate, but is antisymmetric, and so in principle isnot excited if the steps in the facing vanes are identical, although atthese wavelengths, achieving truly identical steps may be difficult ifnot impossible to achieve. Also, it is noted that the graph shown inFIG. 18 does not include the iterative process discussed above, whichshould make the reflected power zero at 170 GHz in the sapphire, butwhich would make the dimension b even larger. The less than 1% reflectedpower in the dielectric is really very acceptable, however, so such aniteration is unnecessary. Nevertheless, the issue of the effect of anevanescent mode is a concern. (A more detailed analysis might provideguidance on whether some small change in the transformer length couldcompensate for such effects, but it would also be useful to have analternative design that does not require such a large step.) Anotheradvantage to a smaller step in the transformer section is that a largestep constricts the coolant channel, making the manufacture thereof moredifficult.

With the foregoing comments in mind, the proposed solution is to havetwo smaller ratio transformers in series, each of which uses a smallerstep, as shown in FIG. 19. The equivalent circuit (for analysispurposes) of the structure shown in FIG. 19 is shown in FIG. 20.

For the equivalent circuit shown in FIG. 20, Y₀ ', Y₀ are thecharacteristic admittances of the transmission line sections, and Y₁,Y₂, . . . Y₅ are the admittances (i.e., the ratio of the current to thevoltage in this transmission line equivalent circuit) at the indicatedterminals. Starting at the right side of FIG. 20, it is seen that Y₁=G.tbd.Y₀ '√ε; Y₂ =(Y₀ ')² /Y₁ ; Y₃ =(Y₀)² /Y₂ ; Y₄ =(Y₀ ')² /Y₃ ; andY₅ =Y₀)² /Y₄. Thus, ##EQU26## The expression set forth in Eq. (10) isonly true, it should be noted, at the design (or center) frequency forwhich the sections of waveguide (or equivalent transmission line) areexactly λ/4 long, A being the wavelength at the design frequency. If itis desired to have Y₅ be equal to Y₀ ', so there is no reflection at thecircuit input (on the left of FIG. 20), then ##EQU27## As in thepreviously provided examples, if b'=0.022 inches, ε=9.4, thenb=1.323b'=0.0291 inches. This value of b may be compared withb=b'e^(1/4) =1.75b'=0.0385 inches, which is a single section transformer(as shown, e.g., in FIG. 13A).

The correction to the transformer lengths to compensate for thediscontinuity capacitance is now, using Marcuvitz, B/Y₀ '=0.0809, whichgives, from the expressions given above, tanδ=-(B/Y₀ ')/√ε^(1/4)ε/^(1/2) -(B/Y₀)² =-0.107, compared to -0.433 for the single sectiontransformer described above in connection with FIG. 13A. Then l/l₀=(π/2+δ)/(π/2)=0.932. This represents only a -7% change in the lengthsrequired for each of the impedance sections (whether they be highimpedance or low impedance) of the transformer.

Referring next to FIG. 13B, there is shown a partial cross-sectionalview of the distributed window with 2-section transformers 15" to matchthe dielectric strips to parallel waveguide without dielectric, which isin turn matched to free space by the taper sections.

A detailed analysis of the dual transformer structure 15" of FIG. 13B(and FIGS. 19 and 20), including the frequency dependence, is similar tothat which has been presented above relative to the single sectiontransformer 15' of FIG. 13A, but with the addition of a furthertransformer section on each side of the dielectric. When such ananalysis is carried out, the results shown in the graph of FIG. 21 areachieved.

In FIG. 21, for the vertical scale that is used, the reflections ofpower may look rather high. However, they are still much reduced, theuseful bandwidth is increased, compared with the example when nomatching section is used (FIG. 15). More important, the reflection inthe dielectric achieved using the dual-transformer section depicted inFIGS. 13B, 19, 20 and 21, is less than one percent over more than 6 GHz.This means that the dissipation in the dielectric will be reduced toless than 60% of the value without the matching transformers.

Another potentially useful aspect of the geometry illustrated in FIGS.13B and 19, which may improve the transmitted efficiency, is thepossible adjustment available by varying the length of the phase shiftregion 19 (FIG. 13B). The phase shift region (or phase shift section)length can be adjusted as required to adjust the phase relation betweenthe residual reflections due to the taper regions 22 (or taper sections)and the reflections of the region between the tapers (the dielectric andtransformer sections/regions). In particular, the taper reflections maybe put in quadrature with the other reflections, so they do not addconstructively, by making the phase shift regions 19 λ/4 long at thecenter frequency. Alternatively, such regions 19 could be used tosubtract from the residual reflections, although these reflections arevery small at the center frequency (which center frequency is 170 GHz inthe previous examples). The phase shift sections/regions 19 could alsobe used, if the reflections of the dielectric and transformer arenegligible, to ensure that the reflections from the taper at one endcancel those from the other end.

As described above, it is thus seen that the present invention providesa way to widen the bandwidth of a large diameter distributed microwavewindow by using one or more transformer sections as part of the vanestructure of such window.

While the invention has been described with reference to one or twoparticular embodiments thereof, the invention is not intended to be solimited. Numerous variations of the invention could be realized by thoseof skill in the art given the main concepts presented and disclosedherein.

What is claimed is:
 1. A wide bandwidth distributed microwave window foruse within a microwave waveguide (32) comprising:a plurality ofalternating dielectric strips (14) and metallic strips (16) stacked andsealed to form a vacuum barrier (12); said vacuum barrier beingpositioned and sealed so as to provide a physical barrier within theinterior of said waveguide (32); and wherein each of said plurality ofdielectric strips (14) has a substantially rectangular cross-sectionalshape; with a first set of opposing sides being sealed to respectivesides of adjacent ones of said metallic strips (16), and with a secondset of opposing sides fronting the interior of said waveguide, each ofsaid metallic strips (16) has a substantially hexagonal cross-sectionalshape, with a first set of opposing sides being sealed to respectivesides of adjacent ones of said dielectric strips (14), and with a secondand third set of opposing sides of said hexagonal-shaped metallic stripbeing exposed to the interior of said waveguide to form a taper (22),and each of said metallic strips (16) further includes an impedancematching section (15) positioned between the taper (22) and thedielectric strip (14) which comprises at least one quarter wave matchingsection positioned within the window to render the dielectric strip (14)non-resonant.
 2. The wide bandwidth microwave window as set forth inclaim 1 wherein said impedance matching section (15') comprises at leasttwo quarter wave matching sections.
 3. The wide bandwidth microwavewindow as set forth in claim 2 wherein the impedance matching section(15') provides a clear line of sight to the dielectric strip (14)without covering or blocking any portion of the dielectric strip.
 4. Themicrowave window as set forth in claim 1, wherein said metallic stripsand dielectric strips of said vacuum barrier are oriented within saidwaveguide to be perpendicular to a transverse electric field componentof an incident wave of electromagnetic microwave radiation that ispropagating through said waveguide.
 5. The microwave window as set forthin claim 1 wherein a plurality of said metallic strips (16) each includeat least one coolant channel (18) that passes longitudinallytherethrough.
 6. The microwave window as set forth in claim 5 whereinthe second and third set of opposing sides of said hexagonal-shapedmetallic strip combine to form a taper (22) on each side of the vacuumbarrier for each one of said metallic strips (16), each of said tapershaving a ridge (26) that extends the length of said metallic strip; saidridge being a distance L from the impedance matching section (15); saidimpedance matching section having a length of nλ/4, where n is aninteger equal to the number of λ/4 sections included in the theimpedance matching section; and said dielectric strip having a thicknessd; whereby the overall thickness of the vacuum barrier (12) fromridge-to-ridge is 2(L+nλ/4)+d, where λ is the free space wavelength ofthe electromagnetic radiation propagating through said waveguide.
 7. Themicrowave window as set forth in claim 6 wherein each dielectric stripis made from sapphire.
 8. Coupling apparatus for coupling microwavepower between the HE₁₁ mode in a first waveguide to the HE₁₁ mode in asecond waveguide, said apparatus comprising:a vacuum barrier (12)separating said first and second waveguide, said vacuum barrierincluding a plurality of parallel dielectric strips (14), eachdielectric strip being separated from an adjacent dielectric strip by ametallic cooling strip (16), the distance between a center line ofadjacent dielectric strips being approximately a distance h, where h<λ₀,where λ₀ is the free space wavelength associated with the microwavepower being coupled between said first and second waveguide, and furtherwherein the metallic cooling strip includes an impedance matchingsection (15') which comprises at least one quarter wave matching sectionof length λ₀ /4 positioned within the vacuum barrier to render thedielectric strip (14) non resonant, the thickness of the vacuum barrierthus being a distance d through said dielectric strips, and a distanced+2(L+nλ₀ /4) through the thickest part of said metallic cooling strips,where L is the distance between a ridge of the metallic cooling strip(16) and the impedance matching section (15'), and n is an integer equalto the number of quarter wave matching sections, whereby each metalliccooling strip extends perpendicularly out from a plane surface of saiddielectric strips a distance L+nλ₀ /4; the dielectric strips of saidvacuum barrier being oriented so as to be longitudinally perpendicularto an electric field component of said microwave power.
 9. A method offorming a vacuum barrier that separates first and second waveguides,said method comprising the steps of:(a) forming a plurality ofdielectric strips so that the thickness of said vacuum barrier is adistance d through said dielectric strips; (b) forming a plurality ofmetallic cooling strips so that there is a distance d+2(L+nλ₀ /4)through the thickest part of said metallic cooling strips, each metalliccooling strip extending perpendicularly out from a plane surface of saiddielectric strips a distance L+nλ₀ /4, where n is an integer, L is thedistance between a ridge of the metallic cooling strip and a quarterwave matching section, and λ₀ is the free space wavelength associatedwith microwave power being transmitted through the waveguides, andwherein the metallic cooling strips include at least one quarter wavematching section positioned within the window to render the dielectricstrip non-resonant; (c) adjoining a cooling strip on each side of eachdielectric strip, thereby forming a barrier, such that the distancebetween a center line of adjacent dielectric strips is a distance h,where h<λ₀ ; and (d) mounting said barrier between said first and secondwaveguides so as to separate said first and second waveguides, andorienting the dielectric strips to be perpendicular to an electric fieldcomponent of microwave power propagating through said first and secondwaveguides.
 10. The method of claim 9 wherein step (b) includes formingthe metallic cooling strips to include at least two quarter wavematching sections positioned within the window.